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mic24053 12v, 9a high - efficiency buck regulator superswitcher ii ? hyper speed control, superswitcher ii, and any capacitor are trademarks of micrel, inc. hyper light load is a registered trademark of micrel, inc. micr el inc. ? 2180 fortune drive ? san jose, ca 95131 ? usa ? tel +1 ( 408 ) 944 - 0800 ? fax + 1 (408) 474 - 1000 ? http://www.micrel.com november 2012 m9999 -1107 12 -a general description the micrel mic24053 is a constant - frequency, synchronous buck regulator featuring a unique adaptive on - time control architecture. the mic24053 operates over an input supply range of 4.5v to 19v and provides a regulated output of up to 9 a of output current. the output voltage is adjustable down to 0.8v with a guaranteed accuracy of 1%, and the device operates at a switching frequency of 600khz. micrel?s hyper speed control ? architecture allows for ultra - fast transient response while red ucing the output capacitance and also makes (high v in )/(low v out ) operation possible. this adaptive t on ripple control architecture combines the advantages of fixed - frequency operation and fast transient response in a single device. the mic24053 offers a f ull suite of features to ensure protection of the ic during fault conditions. these include undervoltage lockout to ensure proper operation under power - sag conditions, internal soft - start to reduce inrush current, foldback current limit, ? hiccup mode ? shor t - circuit protection , and thermal shutdown. an open - drain power good (pg) pin is provided. the 9a hyper light load ? part, mic24054, is also available on micrel?s web site. all support documentation is available on micrel?s web site at: www.micrel.com . features ? hyper speed control architecture enables - high delta v operation (v in = 19v and v out = 0.8v) - small output capacitance ? 4.5v to 19v voltage input ? 9a output current capability, up to 95% efficiency ? adjustable out put from 0.8v to 5.5v ? 1% feedback accuracy ? any capacitor ? s table - z ero - to - high esr ? 600khz switching frequency ? no external compensation ? power good (pg) output ? fold back current - limit and ?hiccup mode ? short - circuit protection ? supports safe startup into a pre - biased load ? ? 40 c to +125 c junction temperature range ? available in 28- pin 5mm 6mm qfn package applications ? servers, work station s ? routers, switches, and t elecom equipment ? base station s _________________________________________________________________ __________________________________________ typical application efficiency (vin = 12v) vs. output current 50 55 60 65 70 75 80 85 90 95 100 0 2 4 6 8 10 12 output current (a) efficiency (%) 5.0v 3.3v 2.5v 1.8v 1.5v 1.2v 1.0v 0.9v 0.8v vin = 12v
micrel, inc. mic24053 november 2012 2 m9 999 - 110 7 12 - a ordering information part number switching frequency voltage package junction temperature range lead finish MIC24053YJL 600khz adjustable 28 - pin 5mm 6mm qfn ? 40 c to + 125 c pb - free pin configuration 28- pin 5mm 6mm qfn (j l) (top view) pin description pin number pin name pin function 1 pvdd 5v internal linear regulator output . pvdd supply is the power mosfet gate drive supply voltage created by internal ldo from vin. when vin < + 5.5v, pvdd should be tied to pvin pins. a 2.2f ceramic capacitor from the pvdd pin to pgnd (pin 2) must be place d next to the ic. 2, 5, 6, 7, 8, 21 pgnd power ground. pgnd is the ground path for the mic240 53 buck converter power stage. the pgnd pi ns connect to the low - side n - channel internal mosfet gate drive supply ground, the sources of the mosfets, the negative terminals of input capacitors, and the negative t erminals of output capacitors. the loop for the power ground should be as small as poss ible and separate from the signal ground (sgnd) loop. 3 nc no connect. 4, 9, 10, 11, 12 sw switch node output . internal connection for the high - side mosfet source and low - sid e mosfet drain. because of the high - speed switching on this pin, the sw pin shou ld be routed away from sensitive nodes. 13, 14, 15, 16, 17, 18, 19 pvin high - side n - interna l mosfet drain connection input. the pvin operating voltage range is from 4.5v to 19v. input capacitors between the pvin pins and the power ground (pgnd) are requir ed; keep the connection short. 20 bst boost output . bootstrapped voltage to the high - side n - channel mosfet driver. a schottky diode is connected between the pvdd pin and the bst pin. a boost capacitor of 0.1f is connected between the bst pin and the sw p in. adding a small resistor at the bst pin can slow down the turn - on time of high - side n - channel mosfets.
micrel, inc. mic24053 november 2012 3 m9 999 - 110 7 12 - a pin description (continued) pin number pin name pin function 22 cs current sense input . the cs pin senses current by monitoring the voltage across th e low - side mosfet during the off - time. c urrent sensing is necessary for short circuit protection. to sense the current accurately, connect the low - side mosfet drain to sw using a kelvin connection. the cs pin is also the high - side mosfet?s output driver return. 23 sgnd signal ground. sgnd must be connected directly to the ground planes. do not route the sgnd pin to the pgnd pad on the top layer ; see pcb layout recommendations for details. 24 fb feedback input . input to the tr ansconductance amplifier of the control loop. the fb pin is regulated to 0.8v. a resistor divider connecting the feedback to the output is used to adjust the desired output voltage. 25 pg power good output . open - drain o utput. the pg pin is externally tie d with a resistor to vdd. a high output is asserted when v out > 92% of nominal. 26 en enable input . a logic level control of the output. the en pin is cmos - compatible. logic high = enable, logic low = shutdown. in the off state, the device?s supply curren t is gr eatly reduced (typically 5a). do not leave t he en pin floating . 27 vin power supply voltage input . requires bypass capacitor to sgnd. 28 vdd 5v internal linear regulator output . vdd supply is the power mosfet gate drive supply voltage and the sup ply bus for the ic. vdd is created by internal ldo from v in . when v in < + 5.5v, tie vdd to pvin pins. a 1 f ceramic capacitor from the vdd pin to s gnd pins must be place d next to the ic.
micrel, inc. mic24053 november 2012 4 m9 999 - 110 7 12 - a absolute maximum ratings (1) p vin to p gnd ............................................... ? 0.3v to + 2 9 v vin to p gnd ................................................. ? 0.3v to pvin pvdd, vdd to pgnd ..................................... ? 0.3 v to + 6 v v sw , v cs to p gnd ............................. ? 0.3v to ( pvin +0.3v) v bst to v sw ........................................................ ? 0.3v to 6v v bst to p gnd .................................................. ? 0.3v to 3 5 v v fb , v pg to pgnd ............................. ? 0.3v to (vdd + 0.3v) v en to p gnd ....................................... ? 0.3v to ( vin +0.3v ) pgnd to s gnd ............................................ ? 0.3v to + 0.3v junction temperature .............................................. +150c storage temperature (t s ) ......................... ? 65 c to +150 c lead temperature (soldering, 10s) ............................ 260c esd rating (2). ................................................ esd sensitive operating ratings (3 ) supply voltage ( pvin , vin ) .............................. 4.5v to 19v pvdd, vdd supply voltage (pvdd, vdd) ..... 4.5v to 5.5 v enable input ( v en ) .................................................. 0v to v i n junction temperature (t j ) ........................ ? 40 c to + 125 c maximum power dissipation ...................................... note 4 package thermal resistance (4) 5mm x 6mm qfn - 28 ( ja ) ................................ 28 c/w electrical characteristics (5 ) pvin = vin = v en = 12v, v bst ? v sw = 5v; t a = 25c, unless noted. bold values ind icate ? 4 0 c t j + 125 c. parameter condition min . typ . max . units power supply input input voltage range (vin , pvin) 4.5 19 v quiescent supply current v fb = 1.5v (non - switching) 730 1500 a shutdown supply current v en = 0v 5 10 a vdd supply vo ltage vdd output voltage vin = 7v to 19v , i dd = 40ma 4.8 5 5.4 v vdd uvlo threshold vdd rising 3.7 4.2 4.5 v vdd uvlo hysteresis 400 mv dropout voltage (vin ? vdd) i dd = 25ma 380 600 mv dc- dc controller output - voltage adjust range (v out ) 0 .8 5.5 v reference feedback reference voltage 0 c t j 85c ( 1.0%) 0.792 0.8 0.808 v ? 40c t j 125c ( 1.5%) 0.788 0.8 0.812 load regulation i out = 0a to 9a (continuous mode) 0.25 % line regulation vin = 4.5v to 19v 0.25 % fb bias curren t v fb = 0.8v 50 na enable control en logic level high 1.8 v en logic level low 0.6 v en bias current v en = 12v 6 30 a oscillator switching frequency (6) v out = 2.5 v 450 600 750 khz maximum duty cycle (7) v fb = 0v 82 % minimum duty cycle v fb = 1.0v 0 % minimum off - time 300 n s
micrel, inc. mic24053 november 2012 5 m9 999 - 110 7 12 - a electrical characteristics (5) (continued) pvin = vin = v en = 12v, v bst ? v sw = 5v; t a = 25c, unless noted. bold values indicate - 40c t j +125c. parameter condition min . typ . max . units soft - start soft - start t ime 3 ms short - circuit protection peak inductor current - limit threshold v fb = 0.8v, t j = 25 c 12.5 14 20 a v fb = 0.8v, t j = 125c 11.25 14 20 a short - circuit current v fb = 0v 8 a internal fets top - mosfet r ds (on) i sw = 3 a 27 m ? bottom - mosfet r ds (on) i sw = 3 a 10.5 m ? sw leakage current v en = 0v 60 a v in leakage current v en = 0v 25 a power good (pg) pg threshold voltage sweep v fb from low to high 85 92 95 %v out pg hysteresis sweep v fb from high to low 5.5 %v out pg delay time sweep v fb from low to high 100 s pg low voltage sweep v fb < 0.9 v nom , i pg = 1ma 70 200 mv thermal protection overt emperature shutdown t j rising 160 c overt emperature shutdown hysteresis 15 c notes: 1. exceeding th e absolute maximum rating can damage the device. 2 . devices are esd sensitive. handling precautions are recommended. human body model, 1.5k ? in series with 100pf. 3. the device is not guaranteed to function outside its operating range. 4. pd (max) = (t j(max) ? t a )/ ja , where ja depends on the printed circuit layout. a 5in 2 , 4 - layer, 0.62?, fr - 4 pcb with 2oz finish copper weight per layer is u sed for the ja . 5 . specification for packaged product only. 6 . measured in test mode. 7. the maximum duty - cycle is limited by the fixed mandatory off - time ( t off ) of typically 30 0 ns.
micrel, inc. mic24053 november 2012 6 m9 999 - 110 7 12 - a typical characteristics vin operating supply current vs. input voltage 0 5 10 15 20 25 4 7 10 13 16 19 input voltage (v) supply current (ma) v out = 1.8v i out = 0a switching vin shutdown current vs. input voltage 0 15 30 45 60 4 7 10 13 16 19 input voltage (v) shutdown current (a) v en = 0v r en = open vdd output voltage vs. input voltage 0 2 4 6 8 10 4 7 10 13 16 19 input voltage (v) vdd voltage (v) v fb = 0.9v i dd = 10ma feedback voltage vs. input voltage 0.792 0.796 0.800 0.804 0.808 4 7 10 13 16 19 input voltage (v) feedback voltage (v) v out = 1.8v i out = 0a total regulation vs. input voltage -0.2% -0.1% 0.0% 0.1% 0.2% 4 7 10 13 16 19 input voltage (v) total regulation (%) v out = 1.8v i out = 0a to 9a output current limit vs. input voltage 0 5 10 15 20 4 7 10 13 16 19 input voltage (v) current limit (a) v out = 1.8v switching frequency vs. input voltage 350 400 450 500 550 600 650 700 4 7 10 13 16 19 input voltage (v) frequency (khz) v out = 1.8v i out = 0a enable input current vs. input voltage 0 4 8 12 16 4 7 10 13 16 19 input voltage (v) en input current (a) v en = vin pg/v ref ratio vs. input voltage 80% 85% 90% 95% 100% 4 7 10 13 16 19 input voltage (v) v pg threshold/v ref (%) v fb = 0.8v
micrel, inc. mic24053 november 2012 7 m9 999 - 110 7 12 - a typical characteristics (continued) vin operating supply current vs. temperature 0.0 5.0 10.0 15.0 20.0 25.0 30.0 -50 -25 0 25 50 75 100 125 temperature (c) supply current (ma) vin = 12v v out = 1.8v i out = 0a switching 0 2 4 6 8 10 12 14 -50 -25 0 25 50 75 100 125 shutdown current (a) temperature ( c) vin shutdown current vs. temperature vin = 12v i out = 0a v en = 0v vdd uvlo threshold vs. temperature 0 1 2 3 4 5 -50 -25 0 25 50 75 100 125 temperature (c) vdd threshold (v) rising falling hyst feedback voltage vs. temperature 0.788 0.792 0.796 0.800 0.804 0.808 -50 -25 0 25 50 75 100 125 temperature (c) feeback voltage (v) vin = 12v v out = 1.8v i out = 0a load regulation vs. temperature -1.0% -0.5% 0.0% 0.5% 1.0% -50 -25 0 25 50 75 100 125 temperature (c) load regulation (%) vin = 12v v out = 1.8v i out =0a to 9a line regulation vs. temperature -0.6% -0.5% -0.4% -0.3% -0.2% -0.1% 0.0% 0.1% 0.2% 0.3% -50 -25 0 25 50 75 100 125 temperature (c) line regulation (%) vin = 4.5v to 19v v out = 1.8v i out = 0a switching frequency vs. temperature 350 400 450 500 550 600 650 700 -50 -25 0 25 50 75 100 125 temperature (c) frequency (khz) vin = 12v v out = 1.8v i out = 0a vdd vs. temperature 2 3 4 5 6 -50 -25 0 25 50 75 100 125 temperature (c) vdd (v) vin = 12v v out = 1.8v i out = 0a output current limit vs. temperature 0 5 10 15 20 -50 -25 0 25 50 75 100 125 temperature (c) current limit (a) vin = 12v v out = 1.8v
micrel, inc. mic24053 november 2012 8 m9 999 - 110 7 12 - a typical characteristics (continued) switching frequency vs. output voltage 250 300 350 400 450 500 550 600 650 700 750 0 0.5 1 1.5 2 2.5 3 3.5 4 4.5 5 output voltage (v) frequency (khz) vin = 12v i out = 0a feedback voltage vs. output current 0.792 0.796 0.800 0.804 0.808 0 1.5 3 4.5 6 7.5 9 output current (a) feedback voltage (v) vin = 12v v out = 1.8v output voltage vs. output current 1.782 1.787 1.791 1.796 1.800 1.805 1.810 1.814 1.819 0 1.5 3 4.5 6 7.5 9 output current (a) output voltage (v) vin = 12v v out = 1.8v line regulation vs. output current -1.0% -0.5% 0.0% 0.5% 1.0% 0 1.5 3 4.5 6 7.5 9 output current (a) line regulation (%) vin = 4.5v to 19v v out = 1.8v switching frequency vs. output current 500 550 600 650 700 0 2 4 6 8 10 output current (a) frequency (khz) vin = 12v v out = 1.8v output voltage (vin = 5v) vs. output current 3.0 3.4 3.8 4.2 4.6 5.0 0 2 4 6 8 10 12 output current (a) output voltage (v) vin = 5v v fb < 0.8v t a 25oc 85oc 125oc efficiency (vin = 5v) vs. output current 50 55 60 65 70 75 80 85 90 95 100 0 2 4 6 8 10 12 output current (a) efficiency (%) vin = 5v 3.3v 2.5v 1.8v 1.5v 1.2v 1.0v 0.9v 0.8v ic power dissipation (vin = 5v) vs. output current 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 0 1.5 3 4.5 6 7.5 9 output current (a) power dissipation (w) vin = 5v v ou t = 3.3v v ou t = 0.8v die temperature* (vin = 5v) vs. output current 0 20 40 60 80 100 0 1.5 3 4.5 6 7.5 9 output current (a) die temperature (c) vin = 5v v out = 1.8v die temperature* : the temperature measurement was taken at the hottest point on the mic24053 case mounted on a 5in 2 , 4 layer, 0.62?, fr - 4 pcb with 2oz finish coppe r weight per layer; see the thermal measurement s section. actual results depend on the size of the pcb, ambient temperature, and proximity to other heat - emitting components.
micrel, inc. mic24053 november 2012 9 m9 999 - 110 7 12 - a typical characteristics (continued) 50 55 60 65 70 75 80 85 90 95 100 0 2 4 6 8 10 12 efficiency (%) output current (a) efficiency (vin = 12v) vs. output current 5.0v 3.3v 2.5v 1.8v 1.5v 1.2v 1.0v 0.9v 0.8v vin = 12v ic power dissipation (vin = 12v) vs. output current 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 0 1.5 3 4.5 6 7.5 9 output current (a) power dissipation (w) vin = 12v v out = 5v v out = 0.8v die temperature* : the temperature measurement was taken at the hottest point on the mic24053 case mounted on a 5in 2 , 4 layer, 0.62?, fr - 4 pcb with 2oz finish copper weight per layer; see the thermal measurements s ect ion. actual results depend on the size of the pcb, ambient temperature , and proximity to other heat - emitting components.
micrel, inc. mic24053 november 2012 10 m9 999 - 110 7 12 - a functional characteristics
micrel, inc. mic24053 november 2012 11 m9 999 - 110 7 12 - a functional characteristics ( c ontinued)
micrel, inc. mic24053 november 2012 12 m9 999 - 110 7 12 - a functional characteristics (continue d)
micrel, inc. mic24053 november 2012 13 m9 999 - 110 7 12 - a functional diagram figure 1 . mic24053 block diagram
micrel, inc. mic24053 november 2012 14 m9 999 - 110 7 12 - a functional description the mic24053 is a n adaptive on - time synchronous s tep - d own dc/ dc regulator with an internal 5v linear regulator and a power good (p g) output. it is designed to operate over a w ide input - voltage range , from 4.5 v to 19v , and provides a regulated output voltage at up to 9 a of output current . it uses a n adaptive on - time control scheme to get a constant switching frequency and to simplify the control compensation. o ver current protection is implemented without using an external sense resistor. t he device includes an internal soft - start function , which reduces the power supply input surge current at start - up by controlling the output voltage rise time. theory of operation the mic24053 operates in a continuous mode , as shown in figure 1 . continuous mode in continuous mode, the mic24053 feedback pin ( fb ) senses t he output voltage through the voltage divi der ( r1 and r2 ) , and compares it to a 0.8v reference voltage ( v ref ) at the error comparator through a low - gain transconductance (g m ) amplifier. if the feedback voltage decreases and the output of t he g m amplifier is below 0.8v, the error comparator trigger s the control logic and generate s an on - time period. the on - time period length is predetermined by the ?fixed t on estimation? circuitry: 600khz v v t in out ed) on(estimat = eq. 1 where v out is the output voltage and v in is the power stage input voltage. at the end of the on - time period, the internal high - side driver turns o f f the high - side mosfet and the low - side driver turn s on the low - side mosfet . in most cases, t he off - time period length depend s on the feedback voltage . when the feedback voltage decreases and th e output of the g m amplifier is below 0.8v, the on - time period is trigger ed and the off - time period ends. if the off - time period de termined by the feedback voltage is less than the minimum off - time ( t off (min) ) , which is about 3 0 0ns , the mic24053 control lo gic applies the t off (min) instead. t off (min) is required to maintain enough energy in the boost capacitor (c bst ) to drive the high - side mosfet . the maximum duty cycle is obtained from the 30 0ns t off (min) : s s off(min) s max t 300ns 1 t t t d ? = ? = eq. 2 where t s = 1/6 00 khz = 1. 66 s. m i crel does not recommend using the mic24053 with a n off - time close to t off (min) during steady - state operation . also, as v out increases, the internal ripple injection increase s and reduce s the line regulation performance. therefore, the maxi mum output voltage of the mic24053 should be limited to 5.5v and the maximum external ripple injection should be limited to 200mv. please r efer to the setting output voltage subsection in application information for more details. the actual on - time and resulting switching frequency vary with the part - to - part variation in the rise and fall time s of the internal mosfets , the output load current, and variations in the v dd voltage . also, t he minimum t on results in a lower switching frequency in high v in to v out ap plications, such as 18 v to 1.0v. the minimum t on measured on the mic24053 evaluation board is about 100 ns. during load transient s , the switching frequency is changed due to the var ying off - time. to illustrate the control loop operation , we will analyze both the steady - state and load transient scenario s . figure 2 shows the mic24053 control - loop timing during steady - state operation . during s teady - state, the g m amplifier senses the feedback voltage ripple to trigger the on - time period. the feedback voltage ripple is proportional to the output voltage ripple and the inductor current ripple. the on - time is predetermined by the t on estimat or . the termination of the off - time is controlled by the feedback voltage . at the valley of the feedback voltage ripple, which occurs when v fb falls below v ref , the off period ends and the next on - time period is triggered through the control logic circuitry.
micrel, inc. mic24053 november 2012 15 m9 999 - 110 7 12 - a figure 2 . mic24053 control loop timing figure 3 shows the operation of the mic24053 during a load transient . the output voltage drops due to the sudden load increas e , which cause s the v fb to be less than v ref . this cause s the error comparator to trigger an on - time period. at the end of th e on - time period, a minimum off - time ( t off (min) ) is generated to charge c bst because the feedback voltage is still below v ref . then, the next on - time pe riod is triggered because of the low feedback voltage . therefore, the switching frequency changes during the load transient , but returns to the nominal fixed frequency after the output has stabilized at the new load current level . because of the varying du ty cycle and switching frequency, the output recovery time is fast and the output voltage deviation is small in the mic24053 converter. figure 3 . mic24053 load transient response unlike true current - mode control, the mic24053 uses the output voltage ripple to trigger an on - time period. the output voltage ripple is proportional to the inductor current ripple if the esr of the output capacitor is large enough . the mic24053 control loop has the advantage of eliminat ing the need f or slope comp ensation. t o meet the stability requirements , t he mic24053 feedback voltage ripple should be in phase with the inductor current ripple and large enough to be sensed by the g m amplifier and the error comparator. the recommended feedback voltag e ripple is 20mv ~100mv . if a low - esr output capacitor is selected, then the feedback voltage ripple may be too small to be sensed by the g m amplifier and the error comparator. also, the output voltage ripple and the feedback voltage ripple are not necessar ily in phase with the inductor current ripple if the esr of the output capacitor is very low. in these cases , ripple injection is required to ensure proper operation . please refer to the ripple injection subsection in application information for more details about the ripple injection technique . v dd regulator the mic24053 provides a 5v regulated output for input voltage vin ranging from 5.5v to 19v . when vin < 5.5v, tie v dd to the pvin pins to bypass the internal linear regulator . soft - start soft - start reduces the power supply input surge current at start - up by controlling the output voltage rise time. the input surge appears while the output capacitor is charged up. a slower output rise time draw s a lower input surge current. the mic24053 implements an internal digital soft - start by making the 0.8v reference voltage ( v ref ) ramp fr om 0 to 100% in about 3 ms with 9.7mv step s . therefore, the output voltage is controlled to increase slo wly by a stair - case v fb ramp. after the soft - start cycl e ends, the related circuitry is disabled to reduce current consumption. v dd must be powered up at the same time or after v in to make the soft - start function correctly. current limit the mic24053 uses the r ds(on) of the internal low - side power mosfet to sense overcurrent conditions. this method reduces cost, board space , and power losses taken by a discrete current sense resistor . the low - side mosfet is used because it displays much lower parasitic osci llations during switching than the high - side mosfet.
micrel, inc. mic24053 november 2012 16 m9 999 - 110 7 12 - a in each switching cycle of the mic24053 converter, the inductor current is sensed by monitoring the low - side mosfet in the off period. if the peak inductor current is greater than 14 a , the mic24053 turn s off the high - side mosfet and a soft - start sequence is trigge re d. this mode of operation is called ?hiccup mode . ? i ts purpose is to protect the down stream load in case of a hard short. the load current - limit threshold has a fold back characteristic related to the feedback voltage a s shown in figure 4 . current limit threshold vs. feedback voltage 0 4 8 12 16 20 0.0 0.2 0.4 0.6 0.8 1.0 feedback voltage (v) current limit threshold (a) figure 4 . mic24053 current - limit foldback characteristic power good (pg) the power good (pg) pin is an open - drain output that indicates logic high when the output is nominally 92% of its steady - state voltage. a pull - up resistor of more than 10k should be connected from pg to vdd. mosfet gate drive the block diagram ( figure 1 ) shows a bootstrap circuit consisting of d1 (a schottky diode is recommended) and c bst . this circuit supplies energy to the high - side drive circuit. capacitor c bst is charged , while the low - side mosfet is on , and the voltage on the sw pin is approximately 0v. when the high - side mosfet driver is turned on, energy from c bst is used to turn the mosfet on. as the high - side mosfet turns on, the voltage on the sw pin increases to approximately v in . diode d1 is reverse biased and c bst floats high while continuing to keep the high - side mosfet on. the bias current of the high - side driver is less than 10ma , so a 0.1f to 1f capacitor is sufficient to hold the gate voltage with minimal droop for the power stroke (high - side switching ) cycle ; that is, bst = 10ma x 1.67s/0.1f = 167 mv. when the low - side mosfet is turned back on, c bst is recharged through d1. a small resistor ( r g ) , which is in series with c bst , can be used to slow down the turn - on time of the high - side n - channel mosfet. the drive voltage is derived from the v dd supply voltage . the nominal low - side gate drive voltage is v dd and the nominal high - side gate drive voltage is approximately v dd ? v diode , where v diode is the voltage drop across d1. an approximate 30ns delay between the high - side and low - side driver transitions is used to prevent current from simultaneously flowing unimpeded through both mosfets.
micrel, inc. mic24053 november 2012 17 m9 999 - 110 7 12 - a application information inductor selection selecting the output inductor requires v alues for inductance, peak, and rms currents. the input and output voltages and the inductance value determine the peak - to - peak inductor ripple current. general ly, higher inductance values are used with higher input voltages. larger peak - to - peak ripple currents increase the power dissipation in the inductor and mosfets. larger output ripple currents also require more output capacitance to smooth out the larger ri pple current. smaller peak - to - peak ripple currents require a larger inductance value and therefore a larger and more expensive inductor. a good compromise between size, loss , and cost is to set the inductor ripple current to be equal to 20% of the maximum output current. the inductance value is c alculated by equation 3 : out(max) sw in(max) out in(max) out i 20% f v ) v (v v l ? = eq. 3 where: f sw = switching frequency, 6 00khz 20% = ratio of ac ripple current to dc output current v in (max) = maximum power stage input voltage the peak - to - peak inductor current ripple is: l f v ) v (v v i sw in(max) out in(max) out l(pp) ? = ? eq. 4 the peak inductor current is equal to the average output current plus one half of the peak - to - peak inductor current ripple. i l(pk) =i out(max) + 0.5 i l(pp) eq. 5 the rms inductor current is used to calculate the i 2 r losses in the inductor. 12 i i i 2 l(pp) 2 out(max) l(rms) + = eq.6 the proper selection of core material and minimizing the winding resistance is required to m aximiz e efficiency. the high - frequency operation of the mic24053 requires the us e of ferrite mate rials for all but the most cost - sensitive applications. lower - cost iron powder cores may be used , but the increase in core loss will reduce the efficiency of the power supply. this is especially noticeable at low output power. the winding resistance decreases efficiency at the higher output current levels. the winding resistance must be minimized although this usually comes at the expense of a larger inductor. the power dissipated in the inductor is equal to the sum of the core and copper l osses. at higher output loads, the core losses are usually insignificant and can be ignored. at lower output currents, the core losses can be a significant contributor. core loss information is usually available from the magnetics vendor. copper loss in th e inductor is calculated by equation 7 : p inductor (cu) = i l(rms) 2 r winding eq. 7 the resistance of the copper wire, r winding , increases with the temperature. the value of the winding resistance used should be at the operating temperature. p winding(ht) = r winding(20 c) (1 + 0.0042 (t h ? t 20 c )) eq. 8 where: t h = temperature of wire under full load t 20c = ambient temperature r winding(20c) = room temperature winding resistance (usually specified by the manufacturer) output capacitor selection the type of the output capacitor is usually determined by its equivalent series resistance ( e sr ). voltage and rms current capability are two other important factors for selecting the output capacitor. recommended ca pacitor types are c eramic , low - esr aluminum electrolytic, os - con , and poscap . the output capacitor?s esr is usually the main cause of the output ripple. it also affects the stability of the control loop.
micrel, inc. mic24053 november 2012 18 m9 999 - 110 7 12 - a the maximum value of esr is calculated by equation 9 : l(pp) out(pp) c i v esr out eq. 9 where: v out(pp) = peak - to - peak output voltage ripple i l(pp) = peak - to - peak inductor current ripple the total output ripple is a combination of the esr and output capacitance. the total ripple is calculated in equation 10 : ( ) 2 c l(pp) 2 sw out l(pp) out(pp) out esr i 8 f c i v + ? ? ? ? ? ? ? ? = eq. 10 w here: d = duty cycle c out = output capacitance value f sw = switching frequency as described in the theory of operation subsection in the functional description section, the mic24053 require s at least 20mv peak - to - peak ripple at the fb pin to make the g m amplifier and the error comparator behav e properly. also, the output voltage ripple should be in phase with the inductor current. therefore , the output voltage ripple caused by the output cap acitor s ? value should be much smaller than the ripple caused by the output capacitor esr. i f low - esr capacitors, such as ceramic capacitors, are used for the output capacitors, a ripple injection method should be applied to provi de enough feedback voltage ripple . please refer to the ripple injection subsection for more details. the voltage rating of the capacitor should be twice the output voltage for a tantalum and 20% greater for aluminum electrolytic or os - con. the o utput capacitor rms current is calculated in equation 11 : 12 i i l(pp) (rms) c out = eq. 11 the power dissipated in the output capacitor is: out out out c 2 (rms) c ) diss(c esr i p = eq. 12 input capacitor selection the input capacitor for the power stage input ( v in ) shoul d be selected for ripple current rating and voltage rating. tantalum input capacitors may fail when subjected to high inrush currents, caused by turning on the input supply. a tantalum input capacitor?s voltage rating should be at least two times the maxim um input voltage to maximize reliability. aluminum electrolytic, os - con, and multilayer polymer film capacitors can handle the higher inrush currents without voltage de - rating. the input voltage ripple primarily depend s on the input capacitor?s esr. the pe ak input current is equal to the peak inductor current, so: v in = i l(pk) esr cin eq. 13 the input capacitor must be rated for the input current ripple. the rms value of the input capacitor current is determined at the maximum output current. assuming the peak - to - peak inductor current ripple is low: d) (1 d i i out(max) cin(rms) ? eq. 14 the power dissipated in the input capacitor is: p diss(cin) = i cin(rms) 2 esr cin eq. 15 ripple injection the v fb ripple re quired for proper operation of the mic24053 g m amplifier and error comparator is 20mv to 100mv . however, the output voltage ripple is generally designed as 1% to 2% of the output voltage. for a low output voltage, such as 1v, the output voltage ripple is only 10mv to 20mv, and the feedback voltage ripple is less than 20mv. if the feedback voltage ripple is so sma ll that the g m amplifier and error comparator can not sense it, then the mic24053 will lose control and the output voltage is not regulated. t o have some amount of v fb ripple, a ripple injection method is applied for low output voltage ripple applications.
micrel, inc. mic24053 november 2012 19 m9 999 - 110 7 12 - a the applications are divided into three situations according to the amount of the feedback voltage ripple: 1. enough ripple at the feedback voltage due to the large esr of the output capacitors. as shown in figure 5 , the converter is stable without any ripple injection . the feedback voltage ripple is: (pp) l c fb(pp) i esr r2 r1 r2 v out + = eq. 16 where i l(pp) is the peak - to - peak value of the inductor current ripple. 2. inadequate ripple at the feedback voltage due to the small esr of the output capacitors. the output voltage ripple is fed into the fb pin through a feedforward capacitor ( c ff ) in this situation, as shown in figure 6 . the typical c ff value is between 1nf and 100nf. with the f eedforward capacitor, the feedback voltage ripple is very close to the output voltage ripple: (pp) l fb(pp) i esr v eq. 17 3. virtually no ripple at the fb pin volta ge due to the very - low esr of the output capacitors. figure 5 . enough ripple at fb figure 6 . inadequate ripple at fb figure 7 . invisible ripple at fb in this situation, the output voltage ripple is less than 20mv. therefore, additional ripple is injected into the fb pin from the switching node ( sw ) using a resistor ( r inj ) and a capacitor ( c inj ) , as shown in figure 7 . the injected ripple is: = sw div in fb(pp) f 1 d) - (1 d k v v eq. 18 r1//r2 r r1//r2 k inj div + = eq. 19 w here : v in = p ower stage input voltage d = duty c ycle f sw = switching frequency = (r1//r2//r inj ) c ff in e quations 18 and 19 , it is assumed that the time constant associated with c ff must be much greater than the switching period: 1 t f 1 sw << = e q. 20 if the voltage divider resistors ( r1 and r2 ) are in the k range, a c ff of 1nf to 100nf ca n easily satisfy the large time constant requirement . also, a 100nf injection capacitor ( c inj ) is used in order to be considered as short for a wide range of t he frequencies.
micrel, inc. mic24053 november 2012 20 m9 999 - 110 7 12 - a the process of sizing the ripple injection resistor and capacitors is: step 1. select c ff to feed all output ripples into the feedback pin and make sure the large time constant assumption is satisfied. typical choice of c ff is 1nf to 100n f if r1 and r2 are in the k range. step 2. select r inj according to the expected feedback v oltage ripple using equation 19 : d) (1 d f v v k sw in fb(pp) div ? = eq. 21 then the value of r inj is calculated as: 1) k 1 ( (r1//r2) r div inj ? = eq. 22 step 3. select c inj as 100nf, which could be co nsidered as short for a wide range of the frequencies. setting output voltage the mic2 4053 requires two resistors to set the output voltage as shown in figure 8 . the output voltage is determined by equation 23 : ) r2 r1 (1 v v fb out + = eq. 23 where v fb = 0.8v. a typical va lue of r1 can be between 3k and 10k. if r1 is too large, it may allow noise to be introduced into the voltage feedback loop. if r1 is too small, it will decrease the efficiency of the power supply, especially at light loads. once r1 is selected, r2 can b e calculated using: fb out fb v v r1 v r2 ? = eq. 24 figure 8 . voltage - divider configuration in addition to the external ripple injection added at the fb pin, internal ripple injection is added at the inverting input of th e comparator inside the mic24053 , as shown in figure 9 . the inverting input voltage ( v inj ) is clamped to 1.2v. as v out increases , the swing of v inj is clamped. the clamped v inj reduces the line regulation because it is reflected as a dc error on the fb terminal. therefore, the maximu m output voltage of the mic24053 should be limited to 5.5v to avoid this problem. figure 9 . internal ripple injection
micrel, inc. mic24053 november 2012 21 m9 999 - 110 7 12 - a thermal measurements it is a good idea to m easur e the ic?s case temperature to make sure it is within its operating limits. although this might seem like a very elementary task, it is easy to get false results. the most common mistake is to use the standard thermal couple that comes with a the rmal meter. this thermal couple wire gauge is large, typically 22 gauge, and behaves like a heatsink, resulting in a lower case measurement. two methods of temperature measurement are to use a smaller thermal couple wire or an infrared thermometer. if a th ermal couple wire is used , it must be constructed of 36 gauge wire , or higher (smaller wire size), to minimize the wire heatsinking effect. in addition, the thermal couple tip must be covered in either thermal grease or thermal glue to make sure that the t hermal couple junction is making good contact with the case of the ic. omega brand thermal couple (5sc - tt - k - 36- 36) is adequate for most applications. wherever possible, an infrared thermometer is recommended. the measurement spot size of most infrared the rmometers is too lar ge for an accurate reading on small form factor ics. however, an ir thermometer from optris has a 1mm spot size, which makes it a good choice for measuring the hottest point on the case. an optional stand makes it easy to hold the beam on the ic for long periods of time.
micrel, inc. mic24053 november 2012 22 m9 999 - 110 7 12 - a pcb layout guideline s note: to minimize emi and output noise, follow these layout recommendations. pcb l ayout is critical to achieve reliable, stable , and efficient performance. a ground plane is required to control emi and minimize the inductance in power, signal , and return paths. follow these guidelines to e nsure proper mic24053 regulator operation: ic ? a 2.2f ceramic capacitor, which is connected to the pvdd pin, must be located right at the ic. the pvdd pin is ve ry noise sensitive and placement of the capacitor is critical. use wide traces to connect to the pvdd and pgnd pins. ? a 1 f ceramic capacitor must be placed right between vdd and the signal ground ( sgnd ) . sgnd must be connected directly to the ground planes . do not route the sgnd pin to the pgnd pad on the top layer. ? place the ic close to the point - of - load (pol). ? use fat traces to route the input and output power lines. ? keep s ignal and power grounds separate and connected at only one location. input capacit or ? place the input capacitor next. ? place the input capacitors on the same side of the board and as close to the ic as possible. ? keep both the pvin p in and pgnd connections short. ? place several vias to the ground plane close to the input capacitor ground te rminal. ? use either x7r or x5r dielectric input capacitors. do not use y5v or z5u type capacitors. ? do not replace the ceramic input capacitor with any other type of capacitor. any type of capacitor can be placed in parallel with the input capacitor. ? if a ta ntalum input capacitor is placed in parallel with the input capacitor, it must be recommended for switching regulator applications and the operating voltage must be derated by 50%. ? in ?hot - plug? applications, a tantalum or electrolytic bypass capacitor mus t be used to limit the overvoltage spike seen on the input supply when power is suddenly applied. inductor ? keep the inductor connection to the switch node ( sw ) short. ? do not route any digital lines underneath or close to the inductor. ? keep the switch n ode ( sw ) away from the feedback (fb) pin. ? connect t he cs pin directly to the sw pin to accurate ly sense the voltage across the low - side mosfet. ? to minimize noise, place a ground plane underneath the inductor. ? the inductor can be placed on the opposite side of the pcb with respect to the ic. it does not matter whether the ic or inductor is on the top or bottom as long as there is enough air flow to keep the power components within their temperature limits. the input and output capacitors must be placed on th e same side of the board as the ic. output capacitor ? use a wide trace to connect the output capacitor ground terminal to the input capacitor ground terminal. ? phase margin change s as the output capacitor value and esr changes. contact the factory if the out put capacitor is different from what is shown in the bom. ? the feedback trace should be separate from the power trace and connected as near as possible to the output capacitor. sensing a long high current load trace can degrade the dc load regulation. optio nal rc snubber ? place the rc snubber on either side of the board and as close to the sw pin as possible.
micrel, inc. mic24053 november 2012 23 m9 999 - 110 7 12 - a evaluation board schematic figure 10 . schematic of mic24053 evaluation board (j11 , r13, r15 are for testing purposes)
micrel, inc. mic24053 november 2012 24 m9 999 - 110 7 12 - a e valuation board schematic (continued) figure 11 . schematic of mic24053 evaluation board (optimized for smallest footprint)
micrel, inc. mic24053 november 2012 25 m9 999 - 110 7 12 - a bill of materials item part number manufacturer description qty . c1 open c2, c3 12103c475kat2a avx (1) 4.7f ceramic capacitor, x7r, size 1210, 25v 2 grm32dr71e475ka61k murata (2) c3225x7r1e475k tdk (3) c13, c15 open c4, c5 12106d107mat2a avx 100f ceramic capacitor, x5r, size 1210, 6.3v 2 grm32er60j107me20l murata c3225x5r0j107m tdk c6, c7, c10 06035c104kat2a avx 0.1f ceramic capacitor, x7r, size 0603, 50v 3 grm188r71h104ka93d murata c1608x7r1h104k tdk c8 0603zc105kat2a avx 1.0f ceramic capacitor, x7r, size 0603, 10v 1 grm188r71a105ka61d murata c1608x7r1a105k tdk c9 0603zd225kat2a avx 2.2f ceramic capacitor, x5r, size 0603, 10v 1 grm188r61a225ke34d murata c1608x5r1a225k tdk c12 06035c472kaz2a avx 4.7nf ceramic capacitor, x7r, size 0603, 50v 1 grm188r71h472k murata c1608x7r1h472k tdk c14 b41851 f7227m epcos ( 4 ) 220f aluminum capacitor, 35 v 1 c11, c16 open d1 sd103aws mcc (5) 40v, 350ma schottky diode. sod323 1 sd103aws -7 diodes inc (6) sd103aws vishay (7) l1 hcf1305 - 2r2 -r cooper bussmann ( 8 ) 2.2 h inductor, 15 a saturation current 1 r1 crcw06032r21fkea vishay dale 2.21 ? resistor, size 0603, 1% 1 r2 crcw06032r00fkea vishay dale 2.00 ? resistor , size 0603, 1% 1 r3 crcw060319k6fkea vishay dale 19.6k ? resistor, size 0603, 1% 1 r4 crcw06032k49fkea vishay dale 2.49k ? resistor, size 0603, 1% 1 r5 crcw060320k0fkea vishay dale 20.0k ? resistor, size 0603, 1% 1 r6, r14, r17 crcw060310k0fkea vishay dale 10.0k ? resistor, size 0603, 1% 3 r7 crcw06034k99fkea vishay dale 4.99k ? resistor, size 0603, 1% 1 r8 crcw06032k87fkea vishay dale 2.87k ? resist or, size 0603, 1% 1 r9 crcw06032k006fkea vishay dale 2.00k ? resistor, size 0603, 1% 1 r10 crcw06031k18fkea vishay dale 1.18k ? resistor, size 0603, 1% 1 r11 crcw0603806rfkea vishay dale 806? resistor, size 0603, 1% 1 r12 crcw0603475rfkea vishay dale 475 ? resistor, size 0603, 1% 1
micrel, inc. mic24053 november 2012 26 m9 999 - 110 7 12 - a bill of materials (continued) item part number manufacturer description qty . r13 crcw06030000fkea vishay dale 0 ? resistor, size 0603, 5% 1 r15 crcw060349r9fkea vishay dale 49.9 ? resistor, size 0603, 1% 1 r16, r18 crcw06031 r21fkea vishay dale 1.21 ? resistor , size 0603, 1% 2 r20 open all reference designators ending with ?a? open u1 MIC24053YJL micrel. inc. ( 9) 12v, 9a high - efficiency buck regulator 1 notes: 1. avx: www.avx.com . 2. murat a : www.murata.com . 3. tdk: www.tdk.com . 4. epcos: www.epcos.com . 5. mcc : www.mccsemi.com . 6. diode inc.: www.diodes.com . 7. vishay: www.vishay.com . 8. cooper bussmann: www.cooperbussmann.com . 9. micrel, inc.: ww w.micrel.com .
micrel, inc. mic24053 november 2012 27 m9 999 - 110 7 12 - a pcb layout recommendations figure 12 . mic24053 evaluation board top layer figure 13 . mic24053 evaluation board mid - layer 1 (ground plane)
micrel, inc. mic24053 november 2012 28 m9 999 - 110 7 12 - a pcb layout recommendations (cont inued) figure 14 . mic24053 evaluation board mid - layer 2 figure 15 . mic24053 evaluation board bottom layer
micrel, inc. mic24053 november 2012 29 m9 999 - 110 7 12 - a package information (1) 28- pin 5mm 6mm qfn (jl ) note: 1. package information is corre ct as of the publication date. for updates and most current information, go to www.micrel.com .
micrel, inc. mic24053 november 2012 30 m9 999 - 110 7 12 - a micrel, inc. 2180 for tune drive san jose, ca 95131 usa tel +1 (408) 944 - 0800 fax +1 (408) 474 - 1000 web http://www.micrel.com micrel makes no representations or warranties with respect to the accuracy or completeness of the inform ation furnished in this data sheet. this information is not intended as a warranty and micrel does not assume responsibility for its use. micrel reserves the right to change circuitry, specifications and descriptions at any time without notice. no licens e, whether express, implied, arising by estoppel or otherwise, to any intellectual property rights is granted by this document. except as provided in micrel?s terms and c onditions of sale for such products, micrel assumes no liability whatsoever, and micrel disclaims any express or implied warranty relating to the sale and/or use of micrel products including liability or warranties relating to fitness for a particular purp ose, merchantability, or infringement of any patent, copyright or other intellectual property right . micrel products are not designed or authorized for use as components in life support appliances, devices or systems where mal function of a product can rea sonably be expected to result in personal injury. life support devices or systems are devices or systems that (a) are intende d for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to res ult in a significant injury to the user. a purchaser?s use or sale of micrel products for use in life support appliances, devices or systems is a purchaser?s own risk a nd purchaser agrees to fully indemnify micrel for any damages resulting from such use or sale. ? 20 12 micrel, incorporated.

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